Beam forming apparatus and method for an array antenna system

ABSTRACT

A joint channel and Direction of Arrival (DOA) estimation apparatus and method simplified to efficiently estimate a channel impulse response associated with a spatially selective transmission channel occurring in a mobile radio channel are provided. To uniformly process all directions, angles associated with a beam are predetermined according to a preset method. This selection calculates a linear system model with regular spatial sampling using regular spatial separation of beam angles. The novel beam forming compensates for a difference between an adaptive array antenna and a sector type antenna using appropriate beam steering according to the calculated linear system model, thereby improving performance and facilitating implementation.

PRIORITY

This application claims priority under 35 U.S.C. § 119(a) to anapplication entitled “Beam Forming Apparatus and Method for an ArrayAntenna System” filed in the Korean Intellectual Property Office on May7, 2004 and assigned Serial No. 2004-32409, the entire contents of whichare incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to an array antenna system. Inparticular, the present invention relates to an apparatus and method foroptimal beam forming for transmitting and receiving high-speed data.

2. Description of the Related Art

Reception quality of radio signals is affected by many naturalphenomena. One of the natural phenomena is temporal dispersion caused bysignals reflected on obstacles in different positions in a propagationpath before the signals arrive at a receiver. With the introduction ofdigital coding in a wireless system, a temporally dispersion signal canbe successfully restored using a Rake receiver or equalizer.

Another phenomenon called fast fading or Rayleigh fading is spatialdispersion caused by signals which are dispersed in a propagation pathby an object located a short distance from a transmitter or a receiver.If signals received through different spaces, i.e., spatial signals, arecombined in an inappropriate phase region, the sum of the receivedsignals is very low in intensity, approaching zero. This becomes a causeof fading dips where received signals substantially disappear, and thefading dips frequently occur when a length corresponds to a wavelength.

A known method of removing fading is to provide an antenna diversitysystem to a receiver. The antenna diversity system includes two or morespatially separated reception antennas. Signals received by therespective antennas have low relation in fading, reducing thepossibility that the two antennas will simultaneously generate thefading dips.

Another phenomenon that significantly affects radio transmission isinterference. The interference is defined as an undesired signalreceived on a desired signal channel. In a cellular radio system, theinterference is directly related to a requirement of communicationcapacity. Because radio spectrum is a limited resource, a radiofrequency band given to a cellular operator should be efficiently used.

Due to the spread of cellular systems, research is being conducted on anarray antenna structure connected to a beam former (BF) as a new schemefor increasing traffic capacity by removing an influence of theinterference and fading. Each antenna element forms a set of antennabeams. A signal transmitted from a transmitter is received by each ofthe antenna beams, and spatial signals experiencing different spatialchannels are maintained by individual angular information. The angularinformation is determined according to a phase difference betweendifferent signals. Direction estimation of a signal source is achievedby demodulating a received signal. A direction of a signal source isalso called a “Direction of Arrival (DOA).”

Estimation of DOAs is used to select an antenna beam for signaltransmission to a desired direction or steer an antenna beam in adirection where a desired signal is received. A beam former estimatessteering vectors and DOAs for simultaneously detected multiple spatialsignals, and determines beam-forming weight vectors from a set of thesteering vectors. The beam-forming weight vectors are used for restoringsignals. Algorithms used for beam forming include Multiple SignalClassification (MUSIC), Estimation of Signal Parameters via RotationalInvariance Techniques (ESPRIT), Weighted Subspace Fitting (WSF), andMethod of Direction Estimation (MODE).

An adaptive beam forming process depends upon having correct informationon spatial channels. Therefore, adaptive beam forming can be acquiredonly after estimation of the spatial channels. This estimation shouldconsider not only temporal dispersion of channels but also DOAs of radiowaves received at a reception antenna.

For estimation of spatial channels, a reception side requires thearrangement of an array antenna having K_(a) antenna elements. Such anarray antenna serves as a spatial low-pass filter having a finitespatial resolution. The term “spatial low-pass filtering” refers to anoperation of dividing an incident wave (or impinging wave) of an arrayantenna into spatial signals that pass through different spatialregions. A receiver having the foregoing array antenna combines a finitenumber, N_(b), of spatial signals, for beam forming. As described above,the best possible beam forming requires information on DOAs and atemporally dispersed channel's impulse response for the DOAs. A value ofthe N_(b) cannot be greater than a value of the K_(a), and thusrepresents the number of resolvable spatial signals. The maximum value,max(N_(b)), of the N_(b) is fixed according to a structure of the arrayantenna.

FIG. 1 illustrates an example of a base station (or a Node B) with anarray antenna, which communicates with a plurality of mobile stations(or user equipments). Referring to FIG. 1, a base station 115 has anarray antenna 110 comprised of 4 antenna elements. The base station 115has 5 users A, B, C, D and E located in its coverage. A receiver 100selects signals from desired users from among the 5 users, by beamforming. Because the array antenna 110 of FIG. 1 has only 4 antennaelements, the receiver 100 restores signals from a maximum of 4 users,e.g., signals from users A, B, D and E as illustrated, by beam forming.

FIG. 2 illustrates spatial characteristics of beam forming for selectinga signal from a user A, by way of example. As illustrated, if a veryhigh weight, or gain, is applied to a signal from a user A, a gainapproximating zero is applied to signals from the other users.

In an antenna diversity system using an array antenna, resolvable beamsare associated with DOAs of max(N_(b)) maximum incident waves. Actually,the total number of incident waves is much greater than max(N_(b)), andis subject to change according to a mobile environment. In order toachieve beam forming, a receiver should acquire information on DOAs, andthe acquisition of DOA information can be achieved through DOAestimation. However, estimated DOAs are not regularly spaced apart fromeach other. Therefore, in a digital receiver, conventional beam formingincludes irregular spatial samplings. A final goal of beam forming is toseparate an incident wave so as to fully use spatial diversity in orderto suppress fading. However, its latent faculty is limited by thestructure of an array antenna having a finite spatial resolution.

In the conventional beam-forming methods, differentiation betweenspatially selective transmission channels for radio mobile communicationincludes three separate steps a first step of estimating spatialchannels, a second step of estimating DOAs based on the estimatedspatial channels, and a third step of estimating a spatial and temporalchannel impulse characteristic for a beam forming algorithm using theestimated spatial channels and the estimated DOAs. This 3-step methodhas a heavy implementation load and causes considerable signalprocessing cost in operation and a lack of robustness due to estimationerrors.

SUMMARY OF THE INVENTION

It is, therefore, an object of the present invention to simply implementanalog and digital front ends of a radio communication system bycalculating a linear system model using regular spatial samplings.

It is another object of the present invention to provide an apparatusand method for efficiently estimating a spatially selective transmissionchannel's impulse response in a mobile radio channel needed fortransmitting transmission data at a possible minimum bit error rate(BER) or with possible maximum throughput.

According to one aspect of the present invention, there is provided abeam forming apparatus for an antenna diversity system with an arrayantenna having a plurality of antenna elements. The apparatus comprisesan interference and noise calculator for estimating interference power R_(DOA) and spectral noise density N₀ of a radio channel using a signalreceived through the radio channel, and calculating total noise power ofthe radio channel according to the interference power and the spectralnoise density; a channel estimator for calculating a directional channelimpulse response matrix corresponding to a predetermined number ofdirection-of-arrival (DOA) values using the total noise power, andcombining a phase matrix comprising phase factors associated with theDOA values with the directional channel impulse response matrix tocalculate a combined channel impulse response; and a beam former forperforming beam forming for transmission and reception through the arrayantenna using the combined channel impulse response.

According to another aspect of the present invention, there is provideda beam forming method for an antenna diversity system with an arrayantenna having a plurality of antenna elements. The method comprises thesteps of estimating interference power R _(DOA) and spectral noisedensity N₀ of a radio channel using a signal received through the radiochannel, and calculating total noise power of the radio channelaccording to the interference power and the spectral noise density;calculating a directional channel impulse response matrix correspondingto a predetermined number of direction-of-arrival (DOA) values using thetotal noise power, and combining a phase matrix including phase factorsassociated with the DOA values with the directional channel impulseresponse matrix to calculate a combined channel impulse response; andperforming beam forming for transmission and reception through the arrayantenna using the combined channel impulse response.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other objects, features and advantages of the presentinvention will become more apparent from the following detaileddescription when taken in conjunction with the accompanying drawings inwhich:

FIG. 1 illustrates an example of a conventional base station with anarray antenna, which communicates with a plurality of mobile stations;

FIG. 2 is a polar plot illustrating conventional spatial characteristicsof beam forming for selecting a signal from one user;

FIG. 3 is a polar plot illustrating spatial characteristics for theconventional null steering beam forming;

FIG. 4 is a polar plot illustrating spatial characteristics for theconventional beam forming that performs irregular spatial matchingfiltering, which maximizes received energy in the case of additive whiteGaussian noise (AWGN);

FIG. 5 is a block diagram illustrating a structure of a receiver in anarray antenna system according to an embodiment of the presentinvention;

FIG. 6 is a flowchart illustrating a joint channel and DOA estimationoperation according to an embodiment of the present invention;

FIG. 7 is a diagram for a description of a synthetic scenario for an8-path test channel according to an embodiment of the present invention;

FIG. 8 is a polar plot illustrating spatial characteristics for regularspatial sampling according to an embodiment of the present invention;

FIG. 9 is a diagram illustrating performance acquired for the syntheticscenario of the 8-path test channel according to an embodiment of thepresent invention; and

FIG. 10 is a flowchart illustrating a beam forming operation accordingto another embodiment of the present invention.

DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS

Several embodiments of the present invention will now be described indetail with reference to the accompanying drawings. In the followingdescription, a detailed description of known functions andconfigurations incorporated herein has been omitted for conciseness.

The present invention described below does not consider DOAs of maximumincident waves that need irregular spatial sampling, in performing beamforming by estimating spatial channels in an antenna diversity system.The irregular spatial sampling requires accurate time measurement andtime-varying reconstruction filtering, and is more complex than aregular sampling strategy in implementation. Therefore, the presentinvention pre-calculates a linear system model beginning at regularspatial sampling that uses regular spatial separation for a beam angle,thereby dramatically the reducing complexity of channel estimation.

The conventional beam forming comprises a first step of estimatingspatial channels, a second step of estimating DOAs based on theestimated spatial channels, and a third step of estimating a spatial andtemporal channel impulse characteristic for a beam forming algorithmusing the estimated spatial channels and the estimated DOAs. Incontrast, the present invention simplifies the beam forming with onlythe third step by removing the first and second steps that require alarge number of calculations. Therefore, spatial and temporal channeland DOA estimation that uses, for example, a maximum likelihood (ML)selection scheme is possible.

A system model applied to the present invention will first be described.

A burst transmission frame of a radio communication system has burstsincluding two data transport parts (also known as sub-frames) eachcomprised of N data symbols. Mid-ambles which are training sequencespredefined between a transmitter and a receiver, having L_(m) chips, areincluded in each data carrying part so that channel characteristics andinterferences in a radio section can be measured. The radiocommunication system supports multiple access based on TransmitDiversity Code Division Multiple Access (TD-CDMA), and spreads each datasymbol using a Q-chip Orthogonal Variable Spreading Factor (OVSF) codewhich is a user specific CDMA code. In a radio environment, there are Kusers per cell and frequency band, and per time slot. As a whole, thereare K_(i) inter-cell interferences.

A base station (or a Node B) uses an array antenna having K_(a) antennaelements. Assuming that a signal transmitted by a k^(th) user (k=1, . .. , K) is incident upon (impinges on) the array antenna in k_(d) ^((d))different directions, each of the directions is represented by acardinal identifier k_(d) (k_(d)=1, . . . , K_(d) ^((d))). Then, a phasefactor of a k_(d) ^(th) spatial signal which is incident upon the arrayantenna from a k^(th) user (i.e., a user #k) through a k_(a) ^(th)antenna element (i.e., an antenna element k_(a) (k_(a)=1, . . . ,K_(a))) is defined as

$\begin{matrix}{{\Psi\left( {k,k_{a},k_{d}} \right)} = {2\pi\;{\frac{l^{(k_{a})}}{\lambda} \cdot {\cos\left( {\beta^{({k,k_{d}})} - \alpha^{(k_{a})}} \right)}}}} & (1)\end{matrix}$

In Equation (1), α^((k) ^(a) ⁾ denotes an angle between a virtual lineconnecting antenna elements arranged in a predetermined distance fromeach other to a predetermined antenna array reference point and apredetermined reference line passing through the antenna array referencepoint, and its value is previously known to a receiver according to astructure of the array antenna. In addition, β^((k,k) ^(d) ⁾ denotes aDOA in radians, representing a direction of a k_(d) ^(th) spatial signalarriving from a user #k on the basis of the reference line, λ denotes awavelength of a carrier frequency, and l^((k) ^(a) ⁾ denotes a distancebetween a k_(a) ^(th) antenna element and the antenna array referencepoint.

For each DOA β^((k,k) ^(a) ⁾ of a desired signal associated with a user#k, a unique channel impulse response observable by a virtualunidirectional antenna located in the reference point is expressed by adirectional channel impulse response vector of Equation (2) belowrepresenting W path channels.h _(d) ^((k,k) ^(d) ⁾=( h _(d,1) ^(k,k) ^(d) ⁾ , h _(d,2) ^((k,k) ^(d) ⁾, . . . h _(d,W) ^((k,k) ^(d) ⁾)^(T)   (2)where a superscript ‘T’ denotes transpose of a matrix or a vector, andan underline indicates a matrix or a vector.

For each antenna element k_(a), W path channels associated with each ofa total of K users are measured. Using Equation (1) and Equation (2), itis possible to calculate a discrete-time channel impulse responserepresentative of a channel characteristic for an antenna k_(a) for auser #k as shown in Equation (3).

$\begin{matrix}{{{\underset{\_}{h}}^{({k,k_{a}})} = {\left( {{\underset{\_}{h}}_{1}^{({k,k_{a}})},{\underset{\_}{h}}_{2}^{({k,k_{a}})},\ldots\mspace{11mu},{\underset{\_}{h}}_{W}^{({k,k_{a}})}} \right)^{T} = {\sum\limits_{k_{d} = 1}^{K_{d}^{k}}{\exp{\left\{ {j\;{\Psi\left( {k,k_{a},k_{d}} \right)}} \right\} \cdot {\underset{\_}{h}}_{d}^{({k,k_{d}})}}}}}},{k = {1\mspace{11mu}\ldots\mspace{11mu} K}},{k_{a} = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}}} & (3)\end{matrix}$

In Equation (3), h ^((k,k) ^(d) ⁾ denotes a vector representing adiscrete-time channel impulse response characteristic for a k_(d) ^(th)spatial direction, from a user #k. Herein, the vector indicates that thechannel impulse response characteristic comprises directional channelimpulse response characteristics h ₁ ^((k,k) ^(d) ⁾, h ₂ ^((k,k) ^(d) ⁾,. . . , h _(W) ^((k,k) ^(d) ⁾ for W spatial channels. The directionalchannel impulse response characteristics are associated with the DOAsillustrated in Equation (1).

Using a directional channel impulse response vector of Equation (5)below that uses a W×(W·K_(d) ^((k))) phase matrix illustrated inEquation (4) below including a phase factor Ψ associated with a user #kand an antenna element k_(a) and comprises all directional impulseresponse vectors associated with the user #k, Equation (3) is rewrittenas Equation (6).A _(s) ^((k,k) ^(a) ⁾=(e ^(jΨ(k,k) ^(a) ^(,1)) I _(w) ,e ^(jΨ(k,k) ^(a)^(,2)) I _(W) , . . . ,e ^(jΨ(k,k) ^(a) ^(,K) ^(d) ^((k)) ⁾ I _(W)), k=1. . . K, k _(a)=1 . . . K _(a)   (4)where A _(s) ^((k,k) ^(a) ⁾ denotes a phase vector for K_(d) ^((d))directions for a user #k, and I_(w) denotes a W×W identity matrix.h _(d) ^((k))=( h _(d) ^((k,1)T) ,h _(d) ^((k,2)T) , . . . ,h _(d)^((k,K) ^(d) ^((k)) ^()T))^(T) , k=1 . . . K   (5)h ^((k,k) ^(a) ⁾ =A _(s) ^((k,k) ^(a) ⁾ h _(d) ^((k)) , k=1 . . . K, k_(a)=1 . . . K _(a)   (6)

Using a channel impulse response of Equation (6) associated with a user#k, a channel impulse response vector comprised of K·W elements for anantenna element k_(a) for all of K users is written ash ^((k) ^(a) ⁾=(( A _(s) ^((1,k) ^(a) ⁾ h _(d) ⁽¹⁾)^(T),( A _(s) ^((2,k)^(a) ⁾ h _(d) ⁽²⁾)^(T), . . . ,( A _(s) ^((K,k) ^(a) ⁾ h _(d)^((K)))^(T))^(T) , k _(a)=1 . . . K _(a)  (7)

A directional channel impulse response vector having K·W·K_(d) ^((k))elements is defined ash _(d)=( h _(d) ^((1)T) ,h _(d) ^((2)T) , . . . ,h _(d)^((K)T))^(T)  (8)where h _(d) ^((k)) denotes a directional channel impulse responsevector for a user #k.

Equation (9) below expresses a phase matrix A _(s) ^((k) ^(a) ⁾ for allof K users for an antenna element k_(a) as a set of phase matrixes foreach user.

$\begin{matrix}{{{\underset{\_}{A}}_{s}^{(k_{a})} = \begin{bmatrix}{\underset{\_}{A}}_{s}^{({1,k_{a}})} & 0 & A & 0 \\0 & {\underset{\_}{A}}_{s}^{({2,k_{a}})} & A & 0 \\M & M & O & \; \\0 & 0 & A & {\underset{\_}{A}}_{s}^{({K,k_{a}})}\end{bmatrix}},{k_{a} = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}}} & (9)\end{matrix}$

In Equation (9), ‘0’ denotes a W×(W·K_(d) ^((k))) all-zero matrix, andthe phase matrix A _(s) ^((k) ^(a) ⁾ has a size of (K·W)×(K·W·K_(d)^((k))). Then, for Equation (7), a channel impulse response vector forall of K_(d) ^((k)) signals for all of K users at an antenna elementk_(a) can be calculated byh ^((k) ^(a) ⁾ =A _(s) ^((k) ^(a) ⁾ h _(d) , k _(a)=1 . . . K _(a)  (10)

Using Equation (10), a combined channel impulse response vector havingK·W·K_(a) elements is written ash =( h ^((1)T) ,h ^((2)T) , . . . ,h ^((K) ^(a) ^()T))^(T)   (11)

That is, a phase matrix A _(s) for which all of K_(d) ^((k)) spatialsignals for all of K users for all of K_(a) antenna elements are takeninto consideration is defined as Equation (12), and a combined channelimpulse response vector h is calculated by a phase matrix and adirectional channel impulse response vector as shown in Equation (13).A _(s) =A _(s) ^((1)T) ,A _(s) ^((2)T) , . . . ,A _(s) ^((K) ^(a)^()T))^(T)   (12)h=A _(s) h _(d)   (13)

The matrix A _(s), as described above, is calculated using β^((k,k) ^(d)⁾ representative of DOAs for the spatial signals for each user.

The directional channel impulse response vector h _(d) comprises aninfluence of interference power and noises. The possible number ofinterferences incident upon a receiver is expressed asL=L _(m) −W+1   (14)where L_(m) denotes a length of a mid-amble as described above, and Wdenotes the number of path channels.

A user specific mid-amble training sequence known between a transmitterand a receiver is comprised of L×KW Toeplitz matrixes G ^((k))representing mid-amble sequences for a user #k.G =( G ⁽¹⁾ ,G ⁽²⁾ , . . . ,G ^((K)))   (15)

A reception signal vector associated with a mid-amble, received from anantenna element k_(a), is expressed ase ^((k) ^(a) ⁾ =Gh ^((k) ^(a) ⁾ +n ^((k) ^(a) ⁾ ,k _(a)=1 . . . K _(a)  (16)

In Equation (16), n ^((k) ^(a) ⁾ denotes the total noise vectorcomprised of a universal interference and a thermal noise at an antennaelement k_(a), and is comprised of L elements. Then, a K_(a)×L combinednoise vector n representing L noises affecting all of the K_(a) antennaelements is expressed asn =( n ^((1)T) ,n ^((2)T) , . . . ,n ^((K) ^(a) ^()T))^(T)   (17)

Actually, however, only K_(i) interference signals having the highestintensity among a total of L noises are taken into consideration.Herein, K_(i) is predetermined according to the system capacity.Assuming that an angle between the reference line and an incidentdirection estimated for a k_(i) ^(th) interference signal among Kiinterference signals is defined as an incident angle γ^((k) ^(i) ⁾ ofthe corresponding interference signal, a phase factor of a k_(i) ^(th)interference signal incident upon a k_(a) ^(th) antenna element iswritten as

$\begin{matrix}{{{\Phi\left( {k_{i},k_{a}} \right)} = {2\;\pi\;{\frac{l^{(k_{a})}}{\lambda^{*}} \cdot {\cos\left( {\gamma^{(k_{i})} - \alpha^{(k_{a})}} \right)}}}},\mspace{14mu}{k_{i} = {1\mspace{14mu}\ldots\mspace{14mu} K_{i}}},{k_{a} = {1\mspace{14mu}\ldots\mspace{14mu} K_{a}}}} & (18)\end{matrix}$

Assuming that a reception vector associated with an interference signalk_(i) is defined as n _(i) ^((k) ^(i) ⁾, a noise vector n ^((k) ^(a) ⁾for a k_(a) ^(th) antenna element becomes

$\begin{matrix}{{{\underset{\_}{n}}^{(k_{a})} = {{\sum\limits_{k_{i} = 1}^{k_{i}}{{\mathbb{e}}^{{j\Phi}{({k_{i},k_{a}})}}{\underset{\_}{n}}_{i}^{(k_{i})}}} + {\underset{\_}{n}}_{th}^{(k_{a})}}},{k_{a} = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}}} & (19)\end{matrix}$

In Equation (19), a vector n _(th) ^((k) ^(a) ⁾ denotes a thermal noisemeasured at an antenna element k_(a) having a double-sided spectralnoise density N₀/2, a lower-case letter ‘e’ denotes an exponentialfunction of a natural logarithm, and N₀ denotes spectral noise density.

However, because of spectrum forming by modulation and filtering, ameasured thermal noise is generally a non-white noise. The non-whitenoise has a thermal noise covariance matrix having a normalized temporalcovariance matrix R _(th) of a colored noise as shown in Equation (20).R _(th) =N ₀ {tilde over (R)} _(th)   (20)

In Equation (20), ‘˜(tilde)’ indicates an estimated value, and adescription thereof will be omitted herein for convenience. If aKronecker symbol shown in Equation (21) below is used, an L×L covariancematrix R _(n) ^((u,v)) meaning noise power between an u^(th) antennaelement and a v^(th) antenna element is written as Equation (22).Herein, u and v each are a natural number between 1 and k_(a).

$\begin{matrix}{\delta_{uv} = \left\{ \begin{matrix}1 & {u = v} \\0 & {else}\end{matrix} \right.} & (21) \\\begin{matrix}{{\underset{\_}{R}}_{n}^{({u,v})} = {E\left\{ {{\underset{\_}{n}}^{(u)}{\underset{\_}{n}}^{{(v)}H}} \right\}}} \\{= {E\left\{ {\left( {{\sum\limits_{k_{i} = 1}^{K_{i}}{{\mathbb{e}}^{{j\Phi}{({k_{i},u})}}{\underset{\_}{n}}_{i}^{(k_{i})}}} + {\underset{\_}{n}}_{th}^{(u)}} \right)\left( {{\sum\limits_{k_{i} = 1}^{K_{i}}{{\mathbb{e}}^{{j\Phi}{({k_{i},v})}}{\underset{\_}{n}}_{i}^{(k_{i})}}} + {\underset{\_}{n}}_{th}^{(v)}} \right)^{H}} \right\}}} \\{= {{E\left\lbrack \left( {\sum\limits_{k_{i} = 1}^{K_{i}}{{\mathbb{e}}^{{{j\Phi}{({k_{i},u})}} - {{j\Phi}{({k_{i},v})}}}{\underset{\_}{n}}_{i}^{(k_{i})}{\underset{\_}{n}}_{i}^{{(k_{i})}H}}} \right) \right\rbrack} + {E\left\{ {{\underset{\_}{n}}_{th}^{(u)}{\underset{\_}{n}}_{th}^{{(v)}H}} \right\}}}} \\{{= {{\sum\limits_{k_{i} = 1}^{K_{i}}{{\mathbb{e}}^{{{j\Phi}{({k_{i},u})}} - {{j\Phi}{({k_{i},v})}}}E\left\{ {{\underset{\_}{n}}_{i}^{(k_{i})}{\underset{\_}{n}}_{i}^{{(k_{i})}H}} \right\}}} + {\delta_{uv}N_{0}{\overset{\sim}{\underset{\_}{R}}}_{th}}}},} \\{u,{v = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}}}\end{matrix} & (22)\end{matrix}$

In Equation (22), E{·} denotes a function for calculating energy, and asuperscript ‘H’ denotes Hermitian transform of a matrix or a vector.Assuming in Equation (22) that interference signals of different antennaelements have no spatial correlation and there is no correlation betweeninterferences and thermal noises, Equation (23) is given. Therefore, inaccordance with Equation (23), energy of a k_(i) ^(th) interferencesignal can be calculated using power of the k_(i) ^(th) interferencesignal.E{n _(i) ^((k) ^(i) ⁾ n _(i) ^((k) ^(i) ^()H)}=(σ^((k) ^(i) ⁾)² ·{tildeover (R)}  (23)

In Equation (23), {σ^((k) ^(i) ⁾)² denotes power of a k_(i) ^(th)interference signal. The L×L normalized temporal covariance matrix R isconstant for all of K_(i) interferences and represents a spectral formof an interference signal, and its value is known to a receiver. The Ris a matrix for calculating correlations between interference signals,and between the interference signals and other interference signals. Thecorrelations are determined according to whether relationship betweenthe interference signals are independent or dependent. If there is ahigh probability that when one interference signal A occurs anotherinterference signal B will occur, a correlation between the twointerference signals is high. In contrast, if there is no relationshipbetween generations of the two interference signals, a correlationbetween the two interference signals is low. Therefore, if there is nocorrelation between interference signals, i.e., if the interferencesignals are independent, {tilde over (R)} has a form of a unit matrix inwhich all elements except diagonal elements are 0s. That is, R _(th) andR are approximately equal to each other as shown in Equation (24) below.{tilde over (R)}≈{tilde over (R)} _(th) ≈I _(L)   (24)

In Equation (24), I_(L) denotes an L×L identity matrix. Thus, Equation(22) can be simplified as

$\begin{matrix}\begin{matrix}{{\underset{\_}{R}}_{n}^{({u,v})} = {{\overset{\sim}{\underset{\_}{R}} \cdot {\sum\limits_{k_{i} = 1}^{K_{i}}{\left( \sigma^{(k_{i})} \right)^{2}{\mathbb{e}}^{{{j\Phi}{({k_{i},u})}} - {{j\Phi}{({k_{i},v})}}}}}} + {\delta_{uv}N_{0}{\underset{\_}{\overset{\sim}{R}}}_{th}}}} \\{= {{{\underset{\_}{r}}_{u,v}\overset{\sim}{\underset{\_}{R}}} + {\delta_{uv}N_{0}{\overset{\sim}{\underset{\_}{R}}}_{th}}}} \\{{\approx {\left( {{\underset{\_}{r}}_{u,v} + {\delta_{uv}N_{0}}} \right)I_{L}}},u,{v = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}}}\end{matrix} & (25)\end{matrix}$

A vector r _(u,v) is an interference signal between an antenna element‘u’ and an antenna element ‘v’, defined by Equation (25) itself.

Using Equation (25), an LK_(a)×LK_(a) covariance matrix of a combinednoise vector n defined in Equation (17) is expressed as

$\begin{matrix}\begin{matrix}{{\underset{\_}{R}}_{n} = {{\begin{bmatrix}{\underset{\_}{r}}_{1,1} & {\underset{\_}{r}}_{1,2} & \ldots & {\underset{\_}{r}}_{1,K_{a}} \\{\underset{\_}{r}}_{2,1} & {\underset{\_}{r}}_{2,2} & \ldots & {\underset{\_}{r}}_{2,K_{a}} \\\vdots & \vdots & ⋰ & \vdots \\{\underset{\_}{r}}_{K_{a},1} & {\underset{\_}{r}}_{K_{a},2} & \ldots & {\underset{\_}{r}}_{K_{a},K_{a}}\end{bmatrix} \otimes \underset{\_}{\overset{\sim}{R}}} + {N_{0}{I_{K_{a}} \otimes {\overset{\sim}{\underset{\_}{R}}}_{th}}}}} \\{= {{{\underset{\_}{R}}_{DOA} \otimes \overset{\sim}{\underset{\_}{R}}} + {N_{0}{I_{K_{a}} \otimes {\overset{\sim}{\underset{\_}{R}}}_{th}}}}} \\{\approx {\left\lbrack {{\underset{\_}{R}}_{DOA} + {N_{0}I_{K_{a}}}} \right\rbrack \otimes \underset{\_}{\overset{\sim}{R}}}} \\{\approx {\left\lbrack {{\underset{\_}{R}}_{DOA} + {N_{0}I_{K_{a}}}} \right\rbrack \otimes I_{L}}}\end{matrix} & (26)\end{matrix}$

In Equation (26), a matrix n denotes interference power, and is definedby Equation (26) itself. Because the matrix n is substantially equal tothe vector r _(u,v), R _(DOA) becomes a Hermitian matrix in whichdiagonal elements are equal to each other. Therefore, if only the upperand lower triangular elements of R _(DOA) are estimated, all of theremaining elements can be determined.

According to Equation (25) and Equation (26), it can be understood thata K_(a)×K_(a) matrix R _(DOA) is associated with only DOAs andinterference power of K_(i) interferences. Assuming herein that there isno spatial correlation between interference signals of different antennaelements, an interference signal between the different antenna elementsbecomes 0. Therefore, R _(DOA) can be determined using only the k_(i)^(th) interference power (σ^((k) ^(i) ⁾)² and the spectral noise densityN₀, and the total noise power R _(n) is calculated by the R _(DOA).

A spatial channel impulse response vector h _(d) is estimated byEquation (27) using the total noise power R _(n) calculated by Equation(26).{tilde over (h)} _(d)=( A _(s) ^(H)(I _(K) _(a) {circle around (×)}G^(H)) R _(n) ⁻¹(I _(K) _(a) {circle around (×)}G ^(H)) A _(s))⁻¹ A _(s)^(H)(I _(k) _(a) {circle around (×)}G ^(H))R _(n) ⁻¹ e  (27)where ‘^(hat)’ denotes an estimated value.

As a result, the total combined reception vector of an array antennahaving LK_(a) antenna elements is

$\begin{matrix}\begin{matrix}{\underset{\_}{e} = \left( {{\underset{\_}{e}}^{{(1)}T},{\underset{\_}{e}}^{{(2)}T},\ldots\mspace{11mu},{\underset{\_}{e}}^{{(K_{a})}T}} \right)^{T}} \\{= \begin{pmatrix}{\underset{\_}{e}}^{(1)} \\{\underset{\_}{e}}^{(2)} \\M \\{\underset{\_}{e}}^{(K_{a})}\end{pmatrix}} \\{= {\begin{pmatrix}{\underset{\_}{Gh}}^{(1)} \\{\underset{\_}{Gh}}^{(2)} \\M \\{\underset{\_}{Gh}}^{(K_{a})}\end{pmatrix} + \begin{pmatrix}{\underset{\_}{n}}^{(1)} \\{\underset{\_}{n}}^{(2)} \\M \\{\underset{\_}{n}}^{(K_{a})}\end{pmatrix}}} \\{= {{\left( {I_{K_{a}} \otimes \underset{\_}{G}} \right)\underset{\_}{h}} + \underset{\_}{n}}} \\{\underset{\_}{e} = {{\left( {I_{K_{a}} \otimes \underset{\_}{G}} \right){\underset{\_}{A}}_{s}{\underset{\_}{h}}_{d}} + \underset{\_}{n}}}\end{matrix} & (28)\end{matrix}$

Finally, a receiver selects a row having the maximum size, i.e., asignal of a corresponding antenna element, from the combined receptionsignal vector of Equation (28).

As described above, multiple calculation processes are needed to acquirea designed signal through beam forming. Among the processes, DOAestimation has the larger proportion. The receiver evaluates signalcharacteristics for all directions of 0 to 360° each time, and regards adirection having a peak value as a DOA. Because this process requires somany calculations, research is being performed on several schemes forsimplifying the DOA estimation.

An effect of beam forming according to estimation performance of DOAswill be described below.

FIG. 3 is a polar plot illustrating spatial characteristics for theconventional null steering beam forming based on a Uniform CircularArray (UCA) having 4 antenna elements and an antenna element spacingd_(element)=λ/2 (where λ denotes a wavelength for the center frequencyof a corresponding frequency band), for 3 incident waves having DOAs of60°, 120° and 240° in the same frequency band. That is, FIG. 3illustrates the case where DOAs of spatial signals are fully known to abeam former and ideal beam forming is possible.

Referring to FIG. 3, solid lines represented by reference numerals 10,12 and 14 indicate incident waves. Assuming that the incident wave 10 ata DOA of 60° is an effective signal and the other two incident waves 12and 14 are interference signals, a first spatial characteristicrepresented by reference numeral 10 a is acquired and a gain of the60°-incident wave 10 is represented by reference numeral 10 b. Assumingthat the incident wave 12 at a DOA of 120° is an effective signal andthe other two incident waves 10 and 14 are interference signals, asecond spatial characteristic represented by reference numeral 12 a isacquired and a gain of the 120°-incident wave 12 is represented byreference numeral 12 b. Finally, assuming that the incident wave 14 at aDOA of 240° is an effective signal and the other two incident waves 10and 12 are interference signals, a third spatial characteristicrepresented by reference numeral 14 a is acquired and a gain of the240°-incident wave 14 is represented by reference numeral 14 b.

As described above, the null steering beam forming provides spatialdiversity as it can separate 3 incident waves. This is possible on theassumption that DOA estimation is perfect and beam forming perfectlyreceives only individual signals. However, in an actual scenario whereseveral hundreds of incident waves are used, the perfect DOA estimationis impossible and effective energy is reduced due to interferencesignals, making it difficult to obtain advantages of the null steeringbeam forming design. In particular, perfect separation of incident wavesis actually impossible.

FIG. 4 is a polar plot illustrating spatial characteristics for theconventional beam forming that performs spatial matching filtering,which maximizes received energy in the case of additive white Gaussiannoise (AWGN), based on the UCA antenna having 4 antenna elements and anantenna element spacing d_(element)=λ/2, for 3 incident waves havingDOAs of 60°, 120° and 240° in the same frequency band.

FIG. 4 illustrates only a spatial characteristic 20 a for a 60°-incidentwave 20, and its gain is represented by reference numeral 20 b. Thespatial characteristic 20 a provides gains 20 c and 20 d not only forthe 60°-incident wave 20 but also the other incident waves 22 and 24. Asdescribed above, the actual beam forming cannot perfectly separate 3incident waves. When DOAs estimated in the receiver are perfectivelyaccurate, there is mutual spatial interference. Fortunately, however,the effective energy remains.

Therefore, an embodiment of the present invention provides a novelsimplified joint spatial and temporal channel and DOA estimation scheme.The foregoing conventional solution is considered to develop irregularspatial sampling according to the estimated DOAs. However, as describedabove, the irregular sampling is more complex than the regular samplingin implementation. Therefore, the embodiment of the present inventionreplaces the irregular spatial sampling with the regular samplingtechnique. This is implemented by using a predetermined number of fixedvalues instead of estimating DOAs in beam forming. A basic concept ofthe present invention and its mathematical description will beintroduced below. In addition, the simulation results appropriate forthe spatial scenario will be presented to support implementationpossibility of the proposed method.

An array antenna that forms beams in several directions represented byDOAs can be construed as a spatial low-pass filter that passes thesignals of only a corresponding direction. The minimum spatial samplingfrequency is given by the maximum spatial bandwidth Bs of a beam former.For a single unidirectional antenna, Bs=1/(2π).

If a spatially periodic low-pass filter characteristic is taken intoconsideration using given DOAs, regular spatial sampling with a finitenumber of spatial samples is possible. Essentially, the number of DOAs,representing the number of spatial samples, i.e., the number ofresolvable beams, is given by a fixed value N_(b). Selection of theN_(b) depends upon the array geometry. In the case of the UCA antennawhere antenna elements are arranged on a circular basis, the N_(b) issimply selected such that it should be equal to the number of antennaelements. In the case of another array geometry, i.e., Uniform LinearArray (ULA), the N_(b) is determined by Equation (29) so that thepossible maximum spatial bandwidth determined for all possible scenarioscan be taken into consideration.N _(b)=┌2πB _(s)┐  (29)

In Equation (29), ‘┌·┐’ denotes a ceiling function for calculating themaximum integer not exceeding an input value. For example, assuming thatthe possible maximum spatial bandwidth is Bs=12/(2π), there are N_(b)=12beams.

In the case where the number of directions, K_(d) ^((k)) (k=1, . . . ,K), is fixed and the regular spatial sampling is implemented accordingto the present invention, the number K_(d) ^((k)) of directions is equalto the number N_(b) of DOAs. Accordingly, in the receiver, a wavetransmitted by a user #k affects the antenna array in the N_(b)different directions. As described above, the directions are representedby cardinal identifiers k_(d) (k_(d)=1, . . . , Nb), and angles β^((k,k)^(d) ⁾ associated with DOAs is taken from a finite set defined as

$\begin{matrix}{B = {\left\{ {\beta_{0},{\beta_{0} + \frac{2\pi}{N_{b}}},{\beta_{0} + {2\frac{2\pi}{N_{b}}L\;\beta_{0}} + {\left( {N_{b} - 1} \right)\frac{2\pi}{N_{b}}}}} \right\}.}} & (30)\end{matrix}$

In Equation (30), β₀ denotes a randomly-selected fixed zero phase angle,and is preferably set to a value between 0 and π/N_(b) [radian]. In theforegoing example where N_(b)=12 beams and β₀=0 are used, Equation (30)calculates Equation (31) below corresponding to a set of anglesincluding 0°, 30°, 60°, . . . , 330°.

$\begin{matrix}{B = \left\{ {0,\frac{\pi}{6},{2\frac{\pi}{6}},\ldots\mspace{11mu},{11\frac{\pi}{6}}} \right\}} & (31)\end{matrix}$

When the set B of Equation (30) is selected, the possible differentvalues of β^((k,k) ^(d) ⁾ are the same for all users k=1, . . . , K. Thevalues are previously known to the receiver. Therefore, the receiver nolonger requires the DOA estimation.

Assuming that there are K_(i)=N_(b) interferences, implementation ofangle domain sampling will be described below. Because all the possiblevalues of Equation (30) are acquired by β^((k,k) ^(d) ⁾ and γ^((k) ^(i)⁾, the β^((k,k) ^(d) ⁾ and γ^((k) ^(i) ⁾ are selected by Equation (32)and Equation (33), respectively.

$\begin{matrix}\begin{matrix}{{\beta^{({k,k_{d}})} = {\beta^{(k_{d})} = {\beta_{0} + {2\;\frac{\pi}{N_{b}}\left( {k_{d} - 1} \right)}}}},} & {{k = {1\mspace{11mu}\ldots\mspace{11mu} K}},} & {k_{d} = {1\mspace{11mu}\ldots\mspace{11mu} N_{b}}}\end{matrix} & (32) \\\begin{matrix}{{\gamma^{(k_{i})} = {\beta_{0} + {2\;\frac{\pi}{N_{b}}\left( {k_{i} - 1} \right)}}},} & {k_{i} = {1\mspace{11mu}\ldots\mspace{11mu} N_{b}}}\end{matrix} & (33)\end{matrix}$

From the β^((k,k) ^(d) ⁾ and γ^((k) ^(i) ⁾, a phase factor of a k_(d)^(th) spatial signal which is incident upon a k_(a) ^(th) antennaelement (k_(a)=1, . . . , K_(a)) from a k^(th) user, and a phase factorof a k_(i) ^(th) interference signal which is incident upon the k_(a)^(th) antenna element are simply calculated by Equation (34).

$\begin{matrix}\begin{matrix}{{{\Psi\left( {k,k_{a},k_{d}} \right)} = {{\Psi\left( {k_{a},k_{d}} \right)} = {2\;\pi\;{\frac{l^{(k_{a})}}{\lambda} \cdot {\cos\left( {\beta^{(k_{d})} - \alpha^{(k_{a})}} \right)}}}}},} \\{{{\Phi\left( {k_{i},k_{a}} \right)} = {{\Phi\left( {k_{d},k_{a}} \right)} = {2\;\pi\;{\frac{l^{(k_{a})}}{\lambda} \cdot \cos}\;\left( {\gamma^{(k_{d})} - \alpha^{(k_{a})}} \right)}}},} \\\begin{matrix}{{k_{i} = {k_{d} = {1\mspace{11mu}\ldots\mspace{11mu} N_{b}}}},} & {{k_{a} = {1\mspace{11mu}\ldots\mspace{11mu} K_{a}}},} & {k = {1\mspace{11mu}\ldots\mspace{11mu} K}}\end{matrix}\end{matrix} & (34)\end{matrix}$

Herein, an angle α^((k) ^(a) ⁾ and a distance l^((k) ^(a) ⁾ are fixed bythe geometry of the array antenna.

The number of columns in the A _(s) defined in Equation (12) isK·W·K_(d) ^((k)). However, if Equation (30) and Equation (34) are used,the number of columns is fixed, simplifying signal processing.

FIG. 5 illustrates a structure of a receiver 100 in an array antennasystem according to an embodiment of the present invention, and FIG. 6is a flowchart illustrating operations of the interference and noiseestimator 140, the channel estimator 150 and the beam former 160 in thereceiver 100. An embodiment of the present invention will now bedescribed with reference to FIGS. 5 and 6.

Referring to FIG. 5, an antenna 110 is an array antenna having antennaelements in predetermined array geometry, and receives a plurality ofspatial signals which are incident thereupon through spaces. By way ofexample, it is shown in FIG. 5 that an incident plane wave from only onedirection is received at each of the antenna elements with a differentphase. Each of multipliers 120 multiplies its associated antenna elementby a weight for the corresponding antenna element, determined by thebeam former 160. A data detector 130 performs frequency down-conversion,demodulation, and channel selection on the outputs of the antennaelements, to which the weights were applied, thereby detecting a digitaldata signal.

Referring to FIG. 6, in step 210, the interference and noise estimator140 estimates interference power R _(DOA) and a spectral noise densityN₀ of thermal noise power using data signals provided from the datadetector 130. A noise power R _(n) which is a covariance matrix of acombined noise vector n is calculated using the estimated interferencepower and spectral noise density. At the initial beam forming, becausethere is no data signal received, the interference power R _(DOA) isinitialized to I_(k) to calculate the noise power R _(n).

An example of estimating the spectral noise power density N₀ is asfollows:

1. Switch off all reception antennas.

2. Sample the complex baseband nose signal prevailing at each analogreception branch.

3. Determine the variance of the complex baseband noise sequence. Thevariance is identical to N₀.

Another method is given by measurement of an absolute receivertemperature T. It is found that N₀=Fk_(B)T, where F denotes a linearnoise figure being dependent upon a type of antenna, k_(B) denotesBoltzman's constant and T denotes the absolute receiver temperature.

Next, the interference power is estimated in the following method.Referring to Equation (26) and assuming that there is no correlationbetween interference signals, only the diagonal elements are requiredfor estimation of the R _(DOA). Assuming that K_(i)=K_(d), power (σ^((k)^(i) ⁾)² of a k_(i) ^(th) interference signal can be obviouslydetermined. Therefore, the diagonal elements become the valuesdetermined by normalizing energies for a maximum of Z interferencesignals among K_(i) interference signals arriving at a k_(a) ^(th)antenna element, into Z as shown in Equation (35).

$\begin{matrix}{\left\lbrack {\underset{\_}{\hat{R}}}_{DOA} \right\rbrack_{k_{i},k_{i}} = {\frac{1}{Z}\;{\sum\limits_{z = 1}^{Z}{{{\hat{n}}_{w,d}^{({k_{a},z})} \approx {\left( \sigma^{(k_{i})} \right)^{2} + N_{0}}}}}}} & (35)\end{matrix}$

In Equation (35), n _(w,d) ^((k) ^(a) ^(,z)) is a vector representing az^(th) noise signal estimated at a k_(a) ^(th) antenna, and Z is aninteger selected from a group of values smaller than K_(i). The value Zis given to use a less number Z of interference signals instead ofestimating all of K_(i) interference signals, thereby reducing thenumber of calculations. However, as shown in Equation (35), the diagonalelements of the interference power are determined according to power ofa k_(i) ^(th) interference signal regardless of the Z.

In step 220, the channel estimator 150 calculates a phase matrix A _(s)by a predetermined number N_(b) of DOA values, calculates a directionalchannel impulse response vector h _(d) using Equation (36), andthereafter, calculates a combined channel impulse response vector usingEquation (13).{circumflex over (h)} _(d)=( A _(s) ^(H)(I _(K) _(a) {circle around(×)}G ^(H)) R _(n) ⁻¹(I _(K) _(a) {circle around (×)}G ^(H)) A _(s))⁻¹ A_(s) ^(H)(I _(K) _(a) {circle around (×)}G ^(H))R _(n) ⁻¹ e   (36)

Herein, a matrix G indicating a mid-amble sequence is a value predefinedbetween a transmitter and a receiver, and e is a combined receptionvector calculated by Equation (28).

In step 230, the beam former 160 calculates steering vectors byperforming adaptive beam forming on all directions for each antennaelement using the calculated combined channel impulse response vector h.Thereafter, the beam former 160 performs beam forming on the estimatedDOAs of the incident wave using the combined channel impulse responsevector and the steering vectors.

Performance of the present invention will now be analyzed below. Asynthetic scenario used in the following analysis considers only 8scatters, termed “8-path test channel.” A channel environment for thesynthetic scenario is illustrated in FIG. 7.

Referring to FIG. 7, a base station and a mobile station maintains adistance of 500 m, and a signal from the base station arrives at themobile station passing through 8 paths given by 8 reflecting pointsdenoted by small circles. All of the 8 paths have the same length of 1Km, corresponding to a delay of approximately 3.34 μs. A moving velocityof the mobile station is assumed to be 100 Km/h.

FIG. 8 is a polar plot illustrating spatial characteristics for beamforming where the novel regular spatial sampling is applied to thesynthetic scenario. The mobile station forms 4 regularly spaced beamswith center angles 0°, 90°, 180°and 270°, for an uniform circular array(UCA) antenna with 4 antenna elements and antenna element spacingd_(element)=λ/2 (where λ is a wavelength for the center frequency of acorresponding frequency band) in the same frequency band. Spatialcharacteristics of the beams are denoted by reference numerals 36, 38,40 and 42. DOAs of actually received incident waves are 60°, 120° and240°, and they are denoted by reference numerals 30, 32 and 34.

As illustrated in FIG. 8, it can be noted that the spatialcharacteristics 36, 38, 40 and 42 of the beams formed using the fixedDOAs provide relatively higher gains compared with the actual incidentwaves 30, 32 and 34.

A fading effect occurring in FIG. 8 because of imperfect separation ofthe incident waves can be reduced by increasing a spatial bandwidth of aspatial filter, for example, by increasing the number of antennaelements. The increase in number of antenna elements means an increasein capability of detecting paths in a broadband system.

FIG. 9 illustrates performances on a signal-to-noise ratio e_(b)/n₀acquired for the synthetic scenario of the 8-path test channel,illustrated in FIG. 7. It is assumed in FIG. 9 that a beam former has amaximum spatial bandwidth of Bs=8/(2π)≈0.222°, the number of antennaelements is K_(a)=8 and the number of users is 1. The performanceillustrated herein is the best bit error ratio (BER) that can beacquired using a receiver in spatial channel models of the Rayleighfading environment 50, the conventional technology 52, the noveltechnology 54 used in accordance with an embodiment of the presentinvention, the null steering environment 56 and the AWGN environment 58.

Referring to FIG. 9, the performance of the novel technology 54, whichis associated with the number K_(a) of antenna elements, approaches theperformance of the AWGN environment 58, compared with the performance ofthe conventional technology 52. If K_(a) is granter than or equal to 8,the novel technology is higher by approximately 0.5 dB than theconventional technology in terms of performance.

The foregoing embodiment aims at the optimal combining (maximum ratiocombining) of various beams received at the array antenna. Unlike this,the fixed beam switching uses a set of fixed beams of equal width whichspan the whole 360° plane. In beam switching, only a particular beam ina desired direction rather than two or more beams is switched totransmit the energy of the particular beam to the user. A descriptionwill now be made of another embodiment of the present invention in whichthe foregoing embodiment is applied to the beam switching scenario.

FIG. 10 is a flowchart illustrating a beam forming operation accordingto another embodiment of the present invention. The beam formingoperation described below can be performed by the receiver of FIG. 5.

Referring to FIG. 10, in step 310, the interference and noise estimator140 calculates a covariance matrix representing a noise power usingestimated interference power and spectral noise density N₀. In step 320,the channel estimator 150 calculates a directional channel impulseresponse vector of Equation (36) using the noise power. The directionalchannel impulse response vector, for which all of N_(b) directions forall of K users are taken into consideration, is expressed ash _(d) =( h _(d) ^((1,1)T), h _(d) ^((1,2)T), . . . , h _(d) ^((1,N)^(b) ^()T), . . . , h _(d) ^((K,1)T), h _(d) ^((K,2)T), . . . , h _(d)^((K,N) ^(b) ^()T))^(T)   (37)

In step 330, the channel estimator 150 evaluates channel estimatedvalues for the directional channel impulse response vector of Equation(37) using the energy for each antenna element and each direction, andranks energies ∥h _(d) ^((k,k) ^(d) ⁾∥² of the directional channelimpulse responses estimated in association with each direction k_(d) inorder of their size for each of all the K users.

In step 340, the channel estimator 150 selects only one direction havingthe maximum channel impulse response energy for each user, maintainsonly a channel impulse response energy of the selected direction andsets energies of all other channel impulse responses to zero, forming amodified directional channel impulse response h _(d,mod). The modifieddirectional channel impulse response, together with the phase matrix A_(s) generated using the fixed DOA values, are used for calculating afinal combined channel impulse response h.

In step 350, the beam former 160 detects only one direction for eachuser for each antenna element by performing beam forming using thecalculated combined channel impulse response. That is, for the combinedchannel impulse response, only one direction is taken into considerationfor each user, making it possible to detect only a signal in a desireddirection.

The alternative embodiment illustrated in FIG. 10 reduces the number ofchannel impulse response estimation values taken into consideration forbeam forming for each user, simplifying signal processing.

As can be understood from the foregoing description, the novel beamformer performs regular spatial sampling instead of estimating DOAsneeded for determining weights, thereby omitting the processes neededfor estimating DOAs without considerably deteriorating the beam formingperformance.

While the invention has been shown and described with reference to acertain embodiments thereof, it will be understood by those skilled inthe art that various changes in form and details may be made thereinwithout departing from the spirit and scope of the invention as definedby the appended claims.

1. A beam forming apparatus for an antenna diversity system with anarray antenna having a plurality of antenna elements, the apparatuscomprising: an interference and noise calculator for estimatinginterference power R _(DOA) and spectral noise density N₀ of a radiochannel using a signal received through the radio channel, andcalculating total noise power of the radio channel according to theinterference power and the spectral noise density; a channel estimatorfor calculating a phase matrix including phase factors associated with apredetermined number of direction-of-arrival (DOA) values and adirectional channel impulse response matrix using the total noise power,and calculating a combined channel impulse response by combining thephase matrix with the directional channel impulse response matrix; and abeam former for performing beam forming for transmission and receptionthrough the array antenna using the combined channel impulse response.2. The beam forming apparatus of claim 1, wherein the number of the DOAvalues is set to a maximum integer not exceeding a product (2πB) of apossible maximum spatial bandwidth (B) of the array antenna and a doublecircle ratio (2π).
 3. The beam forming apparatus of claim 1, wherein thenumber of DOA values is equal to the number of the antenna elementscomprising the array antenna when the array antenna has a uniformcircular array (UCA) geometry.
 4. The beam forming apparatus of claim 1,wherein each of the DOA values is determined by$\beta^{(k_{d})} = {\beta_{0} + {\frac{2\;\pi}{N_{b}}\left( {k_{d} - 1} \right)}}$where β^((k) ^(d) ⁾ denotes a DOA of a k_(d) ^(th) signal, β₀ denotes arandomly selected fixed zero-phase angle, N_(b) denotes the number ofthe DOA values, and k_(d) denotes a direction index which is an integerbetween 1 and the N_(b).
 5. The beam forming apparatus of claim 4,wherein the β₀ has a value between 0 and π/N_(b) radian.
 6. The beamforming apparatus of claim 1, wherein the interference power isexpressed with a Hermitian matrix of which diagonal elements are definedin the following equation and the other elements are all zeros (0s),$\left\lbrack {\hat{\underset{\_}{R}}}_{DOA} \right\rbrack_{k_{i}k_{i}} = {\left( \sigma^{(k_{i})} \right)^{2} + N_{0}}$where ‘{circumflex over (R)}_(DOA)’ denotes the interference power, and(σ^(k) ^(i) ⁾)² denotes power of a k_(i) ^(th) interference signal, andN₀ denotes the spectral noise density.
 7. The beam forming apparatus ofclaim 1, wherein the channel estimator evaluates channel estimationvalues of the directional channel impulse response for each directionfor each user, calculates a modified directional channel impulseresponse by maintaining only a direction showing the maximum channelestimation value and setting channel estimation values for all the otherdirections to zero.
 8. The beam forming apparatus of claim 1, whereinthe beam former determines a spatial channel and a DOA of a desiredoptimal signal using the beam forming.
 9. A beam forming method for anantenna diversity system with an array antenna having a plurality ofantenna elements, the method comprising the steps of: estimatinginterference power R _(DOA) and spectral noise density N₀ of a radiochannel using a signal received through the radio channel, andcalculating total noise power of the radio channel according to theinterference power and the spectral noise density; calculating a phasematrix including phase factors associated with a predetermined number ofdirection-of-arrival (DOA) values and a directional channel impulseresponse matrix using the total noise power, and calculating a combinedchannel impulse response by combining the phase matrix with thedirectional channel impulse response matrix; and performing beam formingfor transmission and reception through the array antenna using thecombined channel impulse response.
 10. The beam forming method of claim9, wherein the number of the DOA values is set to a maximum integer notexceeding a product (2πB) of a possible maximum spatial bandwidth (B) ofthe array antenna and a double circle ratio (2π).
 11. The beam formingmethod of claim 9, wherein the number of DOA values is equal to thenumber of the antenna elements comprising the array antenna when thearray antenna has a uniform circular array (UCA) geometry.
 12. The beamforming method of claim 9, wherein each of the DOA values is determinedby$\beta^{(k_{d})} = {\beta_{0} + {\frac{2\;\pi}{N_{b}}\left( {k_{d} - 1} \right)}}$where β^((k) ^(d) ⁾ denotes a DOA of a k_(d) ^(th) signal, β₀ denotes arandomly selected fixed zero-phase angle, N_(b) denotes the number ofthe DOA values, and k_(d) denotes a direction index which is an integerbetween 1 and the N_(b).
 13. The beam forming method of claim 12,wherein the β₀ has a value between 0 and π/N_(b) radian.
 14. The beamforming method of claim 9, wherein the interference power is expressedwith a Hermitian matrix of which diagonal elements are defined in thefollowing equation and the other elements are all zeros (0s),$\left\lbrack {\underset{\_}{\hat{R}}}_{DOA} \right\rbrack_{k_{i},k_{i}} = {\left( \sigma^{(k_{i})} \right)^{2} + N_{0}}$where ‘{circumflex over (R)} _(DOA)’ denotes the interference power, and(σ^((k) ^(i) ⁾)² denotes power of a k_(i) ^(th) interference signal, andN₀ denotes the spectral noise density.
 15. The beam forming method ofclaim 9, wherein the step of calculating a combined channel impulseresponse comprises the step of evaluating channel estimation values ofthe directional channel impulse response for each direction for eachuser, calculating a modified directional channel impulse response bymaintaining only a direction showing the maximum channel estimationvalue and setting channel estimation values for all the other directionsto zero.
 16. The beam forming method of claim 9, further comprising thesteps of determining a spatial channel and a DOA of a desired optimalsignal using the beam forming.